1. Field of the Invention
Embodiments of the invention generally relate to electronics, and in particular, to electronic communication sent differentially over transmission lines.
2. Description of the Related Art
Electronic devices, such as wireless devices, can be susceptible to interference from electromagnetic waves. Such interference is termed electromagnetic interference (EMI) or radio frequency interference (RFI). Electromagnetic wave emissions are typically regulated by a regulatory body, such as the Federal Communications Commission (FCC) for the U.S. and the CISPR for Europe. In addition, there can be several classes of standards for EMI compliance testing. For example, class A applies to industrial environments, class B applies to residential environments, and open-box applies to equipment, such as computer cards, which are intended to be plugged in to another unit.
EMI compliance requirements are seldom overlooked by product engineering in the specification phase, and EMI compliance is often considered only as an afterthought. EMI requirements cannot be waived. EMI compliance difficulties can lead to very expensive last minute shielding solutions or to expensive redesigns. EMI compliance requirements can also be considered beforehand, which typically results in reliance on heavy shielding that can be over designed and expensive, as there is no good tool that accurately predicts levels of EMI emissions appropriate amounts of shielding. In addition, providing shielding is not at option for open-box equipment.
Wired communications techniques offer relatively large throughput at a relatively low cost. By contrast, while optical communications via optical fibers can have a very large throughput and insignificant EMI, the cost of optical networking is relatively high. Also, while wireless communications techniques exist, the bandwidth associated with wireless communications is relatively low compared to wired communications.
In more recent times, high-speed serial data links have taken over from the prior approach, which used a parallel bus with a number of slower speed digital signals. Examples of high-speed serial interfaces currently in use are high speed USB, XAUI, Fiber-channel, Infiniband, serial ATA (SATA), serial attached SCSI (SAS), Gigabit Ethernet, SFP, XFI, and the like. These high-speed interfaces typically start from half a Gigabit per second (Gb/sec) and are now offering more than 10 Gb/sec data rates, with future information rates of over the 25 Gb/s, even 40 Gb/sec on a single high-speed input/output device (HSIO), such as a serializer/deserializer (SERDES). These interfaces use differential signal lines to carry high-speed digital data.
Differential signaling has several advantages over single-ended signaling. A differential signal is carried with two conductors to convey the signal from the transmitter to the receiver. When received at the receiver, the signal is more immune to noise as noise sources tend to affect both conductors carrying the signal in a similar fashion. While a receiver can reject the common mode signal on a pair of signal lines, radiation of the common mode signal can cause EMI.
One conventional way to reduce the radiation of the common mode signal is to apply a common mode filter. A common mode filter eliminates or reduces the common mode signal before it can be radiated by a radiator, such as the signal lines. For series branches, a common mode filter uses magnetic coupling to present a high impedance to a common mode signal and a low impedance to a differential signal. For parallel branches, a common mode filter presents a low impedance to the common mode signal and a high impedance to the differential signal. However, at high frequencies, magnetic coupling in a common mode filter is reduced, which deteriorates the differential signal.
Spread spectrum clocking (SSC) is another technique used to reduce EMI, but SSC increases jitter and is not permitted by certain communications standards. In addition, SSC does not reduce or eliminate discrete common mode frequency spurs, but rather spreads them around. A spectrum analyzer is typically used to measure EMI, and average power passing through a 1 megahertz (MHz) filter is measured. To reduce the average power measured at a particular frequency, spurs are frequency modulated to shift more than +/−0.5 MHz. A spur will then not be present constantly at a particular 1 MHz frequency bin, but will be present only for a fraction of time. The measured EMI power is thus spread over multiple bins. Therefore, SSC can still result in interference to other devices. In addition, SSC techniques complicate the recovery of the data signal, such as a complementary receiver that can lock to a modulated clock, as well as large enough data FIFOs (first in first out) memories to accommodate a large clock variation. For SSC to operate properly, the SSC modulation of the clock signal used for transmission of data should be at a relatively low frequency rate that is typically much lower than the recommended clock recovery bandwidth specified in the applicable standard. Within these limits, then the phase-locked loop (PLL) within the clock recovery circuit in the receiver should be able to follow the modulation of the SSC-modulated clock signal, and jitter of the recovered clock signal should not be increased.
Many of ordinary skill in the art had previously mistakenly believed that the source of EMI from differential signals was the high slew rates of the differential signals themselves. However, as will be shown in connection with FIGS. 1 and 2A-2E, it is the difference between the slew rates for the non-inverted (true) and inverted (false) single-ended signals of the differential signal that generates a common mode signal, which in turn, creates EMI. The non-inverted (true) and the inverted (false) signals are carried by separate electrical conductors, such as wires. Other mismatches, which also generate a common mode signal, will also be described.
Mismatches in rise and fall times are a problem for high speed SERDES transmitters. Rise time and fall time are related to slew rate. Typically, a rise time or a fall time is defined as the time it takes for a signal to rise from 10% to 90% or vice versa. The slew rate relates to the change in voltage per unit time during a rising edge or a falling edge. It should be noted that the slew rate is not constant during a transition. It is difficult to match the rise and fall times of a high-speed driver. Moreover, even when matched at a particular point, a relatively large mismatch can occur over process-voltage-temperature (PVT) variations, resulting in the generation of relatively large AC components for a common mode signal, which in turn radiates energy at the symbol rate and at multiples of the symbol rate. The current trend of operating multiple SERDES output circuits from a single chip with all the output transmitters operating off of the same clock signal, and the trend in increased symbol rate, both further increase the amount of radiated emissions.
Differential signals can radiate EMI in common mode. The common mode radiation, due to the relatively large area encircling the two signal paths carrying the differential signal and ground, can cause failures in compliance with EMI requirements. When radiating EMI, the common mode signal will typically radiate at the symbol rate and at discrete frequencies related to multiples of the symbol rate.
Several distortions of the high speed differential signals can result in common mode signals being generated. The amount of EMI depends much more on the common mode signal auto-correlation level than on the RMS level of the common mode signal as had previously been believed, and for that reason, the radiation at symbol rate frequency and multiples of the symbol rate frequencies are the most severe. FIG. 1 graphically illustrates several independent sources of common mode radiation in connection with waveforms (A) to (E). Of course, combinations of these sources of common mode radiation can be present. In FIG. 1, time is expressed along a horizontal axis, and voltage magnitude is expressed along the vertical axis for each waveform. Solid lines represent the differential signals, and dotted lines represent a resulting common mode due to distortion of the differential signals. Waveforms (A) include a resulting common mode signal due to uneven rising and falling edges from an output driver of the applicable transmitter. Waveforms (B) include a resulting common mode signal due to duty cycle distortion based on a delayed falling or rising edge. Waveforms (C) include a resulting common mode signal due to skew in the differential output signals. Waveforms (D) include a resulting common mode signal due to uneven output levels. Waveforms (E) include a resulting common mode signal due to a difference in low pass filtering between the output signals.
Waveforms (A) illustrate the effects of having uneven rise and fall times for the two differential outputs. As can be observed in waveforms (A) of FIG. 1, the common mode signal peaks have the same polarity from the steady state level. The common mode waveform depends on the distribution of transitions. This results in the alignment of spectra of each individual peak of common mode waveform so the resulting spectrum has strong components at frequencies harmonically related to the symbol rate.
Waveforms (B) illustrate the effects of duty cycle distortion due to delayed rising or falling edges. Duty cycle distortion also results in a common mode signal peaks that have the same polarity from the steady state level. This type of common mode signal results in a spectrum that has strong discrete frequency components at frequencies harmonically related to symbol rate.
Waveforms (C) illustrate the effects of skew in the output signals. In the illustrated example, one of the output signals is delayed with respect to the other due to, for example, a difference in layout. In the illustrated example, the rise and fall times are equal. The common mode waveform has peaks in both directions from the steady state level. The common mode waveform depends on the differential signal, so therefore power spectrum depends on the data signal; however there are typically no dangerous strong discrete frequency components if the data is random or scrambled.
Waveforms (D) illustrate the effects of the two output signals that are not equal in amplitude. The resulting common mode waveform is an attenuated replica of the differential signal waveform, and therefore there are potentially dangerous discrete frequency components in the spectrum even if the data is random or scrambled.
Waveforms (E) illustrated the effects of one of the differential output signals being filtered more than the other. The imbalance in filtering generates a common mode waveform that depends on the data; however, there are typically no dangerous discrete frequency components in the spectrum if the data is random or scrambed.
FIGS. 2A-2E illustrate simulated power spectrums for examples of differential signals the common mode corresponding to waveforms (A) to (E) of FIG. 1. Frequency normalized to the symbol rate is expressed along the horizontal axis. Power in decibels is expressed along a vertical axis. FIGS. 2A-2E illustrate a simulated power spectrum for data and a corresponding common mode given the distortions described earlier in connection with waveforms (A) to (E) of FIG. 1, respectively. In addition, for the case illustrated in connection with FIG. 2A, the common mode has a first spur at −28 decibels relative to the carrier (dBc) and a second spur at −31 dBc. For the case illustrated in connection with FIG. 2A, the common mode has a first spur at −21 dBc and a second spur at −25 dBc.
As illustrated by the various power spectrums of FIGS. 2A-2E, not all types of distortion lead to EMI problems. The worst EMI offenders are typically discrete frequencies (frequency spurs) that rise above other frequencies. Thus, typically, the most dangerous distortions are ones for which peaks from the steady-state common mode value depend on the position of the transition. These distortions are the duty cycle distortion (DCD), which is depicted by waveform (B) of FIG. 1, and the distortion due to uneven rise and fall times depicted by waveform (A) of FIG. 1.
FIG. 3 illustrates coupled inductors for common mode filtering via a common mode choke configuration. The operation of the common mode choke is explained in connection with Equations 1-5. For the common mode choke, the amount of inductance for the inductors L1, L2 should be about the same and is represented by “L” from Equation 3 onwards.
                              V          1                =                ⁢                                            L              1                        ·                                          ⅆ                                  I                  1                                                            ⅆ                t                                              +                      M            ·                                          ⅆ                                  I                  2                                                            ⅆ                t                                                                        (                  Eq          .                                          ⁢          1                )                                          V          2                =                ⁢                              M            ·                                          ⅆ                                  I                  1                                                            ⅆ                t                                              +                                    L              2                        ·                                          ⅆ                                  I                  2                                                            ⅆ                t                                                                        (                  Eq          .                                          ⁢          2                )                                M        =                ⁢                              k            ·                                                            L                  1                                ·                                  L                  2                                                              =                      k            ·            L                                              (                  Eq          .                                          ⁢          3                )                                          V          1                =                ⁢                              (                          1              +              k                        )                    ·          L          ·                                    ⅆ                              I                1                                                    ⅆ              t                                                          (                  Eq          .                                          ⁢          4                )                                          V          2                =                ⁢                              (                          1              +              k                        )                    ·          L          ·                                    ⅆ                              I                2                                                    ⅆ              t                                                          (                  Eq          .                                          ⁢          5                )            
For differential signals, with I1=−I2, the value of the coupling coefficient k is negative. With very tight coupling k=−1, there is virtually no voltage drop on common mode choke for a differential signal. For the common mode, I1=I2 and tight coupling k≈1, so the common mode is attenuated. If the coupling is not very tight, then not only is the common mode not well attenuated, but the differential mode becomes attenuated. For high speed data communication, such as data rates in the range of 10 GHz, the common mode chokes are typically made with bifilar windings wound around a ferrite bead. For these conventional common mode chokes, the absolute value of the coupling factor k is typically in the range between 0.7 and 0.9, which results in a substantial amount of attenuation for the differential mode. In addition, the parasitic capacitance between the windings can cause attenuation of the high frequency components of the signal. The parasitic capacitance across the separate windings can also cause problems. Thus, conventional common mode chokes are typically not useful above about 5 GHz, and the compound use of common mode chokes of high-low-high impedance such as described in U.S. Pat. No. 5,077,543 to Carlile, can become unusable at lower frequencies.
The filtering of common mode spurs on a printed circuit board (PCB) before the signal reaches an effective antenna is another conventional technique. However, the technique is very dependent upon the particular PCB, requires specialized expertise, and can still lead to multiple PCB iterations.
FIGS. 4A and 4B illustrate two cases of different rise and fall slew rates generating a type of common mode signal that has peaks from the steady-state level only in one direction. That type of common mode signal does not depend on the data values, but generates common-mode energy upon transitions with high spurious contents at the symbol rate and multiples of the symbol rate.
FIGS. 4A and 4B illustrate waveforms of pairs of single-ended signals forming differential signals with unbalanced slew rates. In both FIGS. 4A and 4B, voltage is expressed along a vertical axis and time is expressed along a horizontal axis. FIG. 4A illustrates a non-inverted signal 301 and an inverted signal 302 for a differential signal. In the example of FIG. 4A, the non-inverted signal 301 has a higher slew rate for a rising edge than for a falling edge. The inverted signal 302 also has a higher slew rate for a rising edge than for a falling edge. These mismatches in slew rates generate a common mode signal 303 that has only positive pulses (positive peaks above the steady-state level). FIG. 4B illustrates a non-inverted signal 304 that has higher falling edge slew rate than a rising edge slew rate. An inverted signal 305 also has a higher falling edge slew rate than the slew rate of the rising edge. These mismatches generate a common mode signal 306 that has only negative pulses (negative peaks below the steady-state level). The foregoing examples show that common mode signals 303, 306, produced by non-linear common mode generation (FIG. 1, case A and B), result in deterministic waveforms during the transition intervals independent of the randomness of signal data. No matter how much case is placed during design, it will not be possible to prevent common mode signal noise from appearing at boundaries of symbol intervals during the data transition from one state to another. Especially damaging is a resulting common mode signal that is deterministic in nature and repeats at symbol interval boundaries during transitions. The deterministic nature of a common mode signal waveform during transitions causes strong auto-correlation function periodicity at the symbol interval and multiples of the symbol interval, which in turn produces spurs at the symbol rate frequency and multiples of the symbol rate.